Datasheet

r =
'i
L
l
OUT
LM27341, LM27342, LM27341-Q1, LM27342-Q1
SNVS497E –NOVEMBER 2008–REVISED APRIL 2013 www.ti.com
OUTPUT OVERVOLTAGE PROTECTION
The overvoltage comparator turns off the internal power NFET when the FB pin voltage exceeds the internal
reference voltage by 13% (V
FB
> 1.13 * V
REF
). With the power NFET turned off the output voltage will decrease
toward the regulation level.
UNDERVOLTAGE LOCKOUT
Undervoltage lockout (UVLO) prevents the LM27341/LM27342 from operating until the input voltage exceeds
2.75V(typ).
The UVLO threshold has approximately 470 mV of hysteresis, so the part will operate until V
IN
drops below
2.28V(typ). Hysteresis prevents the part from turning off during power up if V
IN
has finite impedance.
THERMAL SHUTDOWN
Thermal shutdown limits total power dissipation by turning off the internal NMOS switch when the IC junction
temperature exceeds 165°C (typ). After thermal shutdown occurs, hysteresis prevents the internal NMOS switch
from turning on until the junction temperature drops to approximately 150°C.
Design Guide
INDUCTOR SELECTION
Inductor selection is critical to the performance of the LM27341/LM27342. The selection of the inductor affects
stability, transient response and efficiency. A key factor in inductor selection is determining the ripple current (Δi
L
)
(see Figure 32).
The ripple current (Δi
L
) is important in many ways.
First, by allowing more ripple current, lower inductance values can be used with a corresponding decrease in
physical dimensions and improved transient response. On the other hand, allowing less ripple current will
increase the maximum achievable load current and reduce the output voltage ripple (see OUTPUT CAPACITOR
section for more details on calculating output voltage ripple). Increasing the maximum load current is achieved by
ensuring that the peak inductor current (I
LPK
) never exceeds the minimum current limit of 2.0A min (LM27341) or
2.5A min (LM27342) .
I
LPK
= I
OUT
+ Δi
L
/ 2 (8)
Secondly, the slope of the ripple current affects the current control loop. The LM27341/LM27342 has a fixed
slope corrective ramp. When the slope of the current ripple becomes significantly less than the converter’s
corrective ramp (see Figure 31), the inductor pole will move from high frequencies to lower frequencies. This
negates one advantage that peak current-mode control has over voltage-mode control, which is, a single low
frequency pole in the power stage of the converter. This can reduce the phase margin, crossover frequency and
potentially cause instability in the converter. Contrarily, when the slope of the ripple current becomes significantly
greater than the converter’s corrective ramp, resonant peaking can occur in the control loop. This can also cause
instability (Sub-Harmonic Oscillation) in the converter. For the power supply designer this means that for lower
switching frequencies the current ripple must be increased to keep the inductor pole well above crossover. It also
means that for higher switching frequencies the current ripple must be decreased to avoid resonant peaking.
With all these factors, how is the desired ripple current selected? The ripple ratio (r) is defined as the ratio of
inductor ripple current (Δi
L
) to output current (I
OUT
), evaluated at maximum load:
(9)
A good compromise between physical size, transient response and efficiency is achieved when we set the ripple
ratio between 0.2 and 0.4. The recommended ripple ratio vs. duty cycle shown below (see Figure 36) is based
upon this compromise and control loop optimizations. Note that this is just a guideline. Please see Application
note AN-1197 SNVA038 for further considerations.
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